Method and apparatus for performing high-density DTMF, MF-R1, MF-R2 detection

ABSTRACT

Detectors determine the presence of valid sinusoids for DTMF, MF-R1 and MF-R2 protocols for encoding dialed digits. The detectors split electrical signals into subbands. Energies within the subbands are analyzed to determine a presence of sinusoids corresponding to frequencies of dialed digits. In one embodiment, the detectors comprise a PS-IIR filter to split the electrical signal into the subbands. The detectors further comprise at least one bank of filters, such as notch filters, corresponding to the number of possible relevant frequencies within the respective subbands. The detectors further comprise detection logic comprising tests, which may include analyzing the output(s) from the bank of filters. Optionally, a preclassifier is employed to predetermine which filters in the banks of filters are to be executed. The detectors, typically deployed in digital signal processors, allow for an increase in the density of detectors and provide robust performance in talk-off situations.

RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.09/696,730, filed Oct. 25, 2000. The entire teachings of the aboveapplication are incorporated herein by reference.

BACKGROUND OF THE INVENTION

The information age has increased the number of users using datacommunication systems. Initially, voice was the primary signal carriedby phone lines. Next, facsimile (i.e., fax) machines became a popularmeans for transferring information, though typically restricted tobusiness environments. Recently, with the advent of the Internet, datacommunications between and among electronic devices has become a commonmode of communications for both businesses and individuals. The increasein mode, user group, and usage has driven the telecommunicationsindustry in general, and service providers in particular, to expandcapacity. One of the limiting factors of service capacity is the size ofdetectors for determining dialed digits by various dialing protocols,such as DTMF.

Detection of the dialed digits at the beginning of a phone connection isas old as the PSTN (Public Switched Telephone Network) itself. Therequirements are well defined and understood. A current interest is inmaximizing the density of DTMF/MF-R1/MF-R2 detection and maintain a highdegree of reliability.

In DTMF (Dual-Tone Multi-Frequency) detection, the dialed digitscorrespond to a row-frequency and a column-frequency, as shown inTable 1. In MF-R1 (Multi-Frequency, One Row) and MF-R2 (Multi-Frequency,Two Rows) detection, two valid frequencies correspond to a dialed digit,although there are no row or column frequencies as in DTMF. Also notethat, in MF-R2, two sets of the frequencies are used depending upon thesignaling direction. The MF-R1 and MF-R2 frequencies are respectivelyshown in Tables 2 and 3. TABLE 1 Hz 1209 1336 1447 1633 697 1 2 3 A 7704 5 6 B 852 7 8 9 C 941 0 0 # D

TABLE 2 Hz 700 900 1100 1300 1500 1700

TABLE 3 FWD 1380 1500 1620 1740 1860 1980 BWD 540 660 780 900 1020 1140

In each case, a bandwidth-test and a twist-test must be passed. In thebandwidth-test, frequency deviations greater than 3.5% must be rejected,and frequency deviations less than 1.5% must be declared as validdigits. In the twist-test, powers at the frequencies of interest must bewithin certain limits with respect to each other. Furthermore, signalsfrom the line that have power levels less than −40 dBm0 must berejected. A condition that is particular to DTMF is falsely detectingdigits when there is speech activity on the line. This condition isknown as talk-off

One way to increase the density of detectors to allow service providersto support more users is to change (i.e., reduce) the sampling rate ofthe incoming analog signal. Changing the sampling rate of the incomingsignal for detection is not a new idea, and there are numerous patentson the subject. However, this concept is poorly applied in many casesand there is a great loss of efficiency. For example, the sampling rateof conversion is often implemented through finite impulse response (FIR)multi-rate filters (MRF), which are inefficient in terms of memory andcomplexity. These are usually implemented by filtering followed by aswitch for straight decimation operations. Therefore, the filtering isperformed at the higher sampling frequency. A better scheme would be thepolyphase implementation of an FIR MRF filter. See P. P. Vaidyanathan,Multi-Rate Systems and Filter Banks, Prentice-Hall, 1993. In thisrepresentation, the switching comes first and the FIR coefficients aredistributed after the switch. In this way, the filtering is performed atthe lower sampling rate.

SUMMARY OF THE INVENTION

Although using FIR MRF filters is a fine idea, it is still inefficientin the context of DTMF due to the aforementioned reasons. For sufficientband isolation, a large number of coefficients might be necessary, and ahigh number of memory locations are necessary to store the filterhistories; thus, a significant computational overhead for filtering isrequired by the FIR-based MRFs. These shortcomings pose serious problemsin high-density applications, since it is desirable to use only theon-chip memory for faster data acquisition and to minimize computationalcomplexity.

In an application, such as telephony, an electrical signal comprisingmultiple sinusoids that encodes dialed digits is split into multiplesubbands. Energies within the subbands are analyzed to determine apresence of sinusoids corresponding to frequencies of dialed digits.

A PS-IR (power-symmetric infinite impulse response) filter may beemployed to split the electrical signal into the subbands. Typically,the electrical signal is split into subbands of 0-1 kHz and 1-2 kHz.Preferably, the PS-IIR filters are implemented in a polyphase form, andall-pass sections composing the PS-IIR filters may be implementedthrough compact realizations. PS-IIR filters are used to maximize thedensity of DTMF/MF-R1/MF-R2 detection and to maintain a high degree ofreliability. PS-IIR filters have ideal features for use in detection ofdialed digits and can be used for all three detector designs. Techniquesemploying PS-IIR filters increase density of detectors and providerobust performance in talk-off situations in DTMF.

The subbands, resulting from whatever band split filter implementationis chosen, are further filtered via a bank of notch filterscorresponding to the number of possible relevant frequencies within therespective subbands; the number of relevant frequencies depends on theencoding protocol, DTMF, MF-R1, or MF-R2. The filters are typicallynotch filters, such as second order infinite impulse response filters.For DTMF detectors, there are four notch filters in the filter banks.For MF-R1 detection, there are two notch filters for the 0-1 kHz subbandand four notch filters for the 1-2 kHz subband. For MF-R2 detection, aforward detector comprises six notch filters in the bank of filters forthe 1-2 kHz subband; a backward detector comprises (i) a notch filter at980 Hz and four other notch filters in the bank of filters for the 0-1kHz subband and (ii) two notch filters in the bank of filters for the1-2 kHz subband. Further, a preclassifier using frequency estimation maybe employed to select only those notch filters corresponding tofrequencies determined to be active in the subbands.

The detectors further comprise detection logic. The detection logicanalyzes the subbands to determine whether the sum of the energiesexceeds a minimum threshold energy level. The detection logic alsoperforms a twist-test to determine whether the energies in the subbandsare within a twist-test threshold. For each subband, the detection logiccompares energy levels between the lowest energy output of the notchfilters and the input energy to the respective bank of filters. Thedetection logic aborts detector execution in the event that an energylevel test determines a discrepancy with a respective specifiedcriterion and reports valid digits after determining the presence ofcorresponding valid sinusoids in the electrical signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, features and advantages of theinvention will be apparent from the following more particulardescription of preferred embodiments of the invention, as illustrated inthe accompanying drawings in which like reference characters refer tothe same parts throughout the different views. The drawings are notnecessarily to scale, emphasis instead being placed upon illustratingthe principles of the invention.

FIG. 1A is a block diagram of an example telephone network havingcentral offices comprising multi-tone detectors designed according tothe principles of the present invention;

FIG. 1B is a block diagram of a media gateway of FIG. 1A employing amulti-tone detector;

FIG. 2 is a block diagram of a circuit board employing the detectors ina central office of FIG. 1A;

FIG. 3 is a block diagram of a DTMF detector executed by a digitalprocessor on a circuit board of FIG. 2;

FIG. 4 is a block diagram of a generic PS-IIR filter executed by theprocessor of and employed by the DTMF detector of FIG. 3;

FIG. 5 is a block diagram of a compact implementation of a first-orderall-pass section employed by the PS-IIR filter of FIG. 4;

FIG. 6 is a plot of frequency responses of (i) a low-pass filter and ahigh-pass filter composing the band-split filter of FIG. 4 and (ii)spectral lines corresponding to frequencies of sinusoids in the DTMFprotocol;

FIG. 7 is a plot of the band-split filter of FIG. 4 and spectral linesof frequencies of sinusoids in the MF-R1 protocol;

FIG. 8 is a plot of the band-split filter of FIG. 4 and spectral linesof frequencies of sinusoids in the MF-R2 protocol;

FIG. 9 is a flow diagram of detection logic for the DTMF protocolemployed by the DTMF detector of FIG. 3;

FIG. 10 is a block diagram of an MF-R1 detector employing the PS-IIRband-split filter of FIG. 4;

FIG. 11A-11C are flow diagrams of processing composing (i) MF-R1detection logic and (ii) MF-R2 forward and backward detection logicemployed by the detector of FIG. 10;

FIG. 12 is a block diagram of a forward detector for the MF-R2 protocolemploying the PS-IIR band-split filter of FIG. 4 and employing thedetection logic of FIGS. 11A-11C;

FIG. 13 is a block diagram of a backward detector for the MF-R2 protocolemploying the PS-IIR band-split filter of FIG. 4 and employing thedetection logic of FIGS. 11A-11C;

FIG. 14 is a block diagram of the DTMF detector of FIG. 3 furthercomprising preclassifiers to improve efficiency; and

FIG. 15 is a flow diagram of an embodiment of the DTMF detection logicof FIG. 9 further comprising preclassifier steps to improve efficiency.

DETAILED DESCRIPTION OF THE INVENTION

A description of preferred embodiments of the invention follows.

With the ever-increasing demand to access the telephony network,switching equipment now has to perform DTMF, MF-R1/R2 detection on manymore channels. High density detection of dialed digits via DTMF, MF-R1,MF-R2 detection is implemented such that density of the number ofchannels per digital signal processor (in a digital implementation) isincreased.

Density in detector channels is increased in two ways. First, bysplitting an input signal into subbands, filters employed by theprocessor operate at slower sampling rates (i.e., lower bandwidths) andare, therefore, less complex. Further, because the filters operate atslower sampling rates, the processor saves instruction cycles for otheroperations, including supporting additional detector channels. Second,density is further improved through the use of a PS-IIR (power symmetricinfinite impulse response) filter in the subband splitter.

The PS-IIR filter can be implemented in a minimal memory form bycascading two first-order all-pass filter sections to implement low- andhigh-pass filters, which compose the PS-IIR filter. In a preferredembodiment, a compact implementation of a first-order all-pass filtersection requires only a single coefficient, a single multiplier, and asingle unit delay (i.e., storage element). Thus, the PS-IIR filterrequires only four storage elements, four coefficients, and very fewprocessor clock cycles.

Banks of notch filters are used to detect the dialed digits encoded insinusoidal signals at various frequencies. The detector may alsocomprise at least one preclassifier that performs frequency estimationon the subband signals to determine frequencies of sinusoids in thesubbands and enables notch filters within the respective banks offilters corresponding to the frequencies of the sinusoids. In this way,notch filters that do not have an affect on processing of the sinusoidsare disabled (i.e., not executed by the processor) to save a significantnumber of clock cycles.

Detection logic executed by the processor comprises several tests beforedeclaring a valid dialed digit. These tests include an energy level testand a twist-test on the energies in the subbands. The detection logicsaves bandwidth in the DSP by exiting upon determining that the inputsignal does not include a sinusoid-encoded, dialed digit. The detectionlogic provides support to minimize talk-off errors. Employing theprinciples of the present invention, it is possible to perform DTMFdetection on 1200 or more channels on a single 250 MHZ TMS320C6202 DSPprocessor. This implementation more than doubles the capacity ofpreceding solutions. Further, this approach is especially robust againstfalse alarms over speech signals due to talk-off

When implemented in a digital form, the detectors can be changed tosupport detection of DTMF, MF-R1, or MF-R2 protocol-encoded dialeddigits. Each detector type is implemented with similar building blocks(e.g., PS-IIR band split filters), so each type has similar size,accuracy, and speed benefits, as discussed above. Differences among thedifferent detector types are found in the detector structures anddetection logic processes to support the various dialed digit protocols,as discussed in detail below in reference to FIGS. 1-15.

FIG. 1A is a block diagram of a data network 100, which is a subset of atelephony system, in which the present invention may be deployed.Customers 110 communicate to central offices 120. The central offices120 communicate to one another via inter-central office transmissions140 via a satellite 130 or via packetized communications across anetwork, such as an IP (Internet Protocol) network 160 or ATM(Asynchronous Transmission Mode) network 170. Access to the IP network160 and ATM network 170 is through a media gateway 150. The principlesof the present invention lend themselves to any system employing tonedetection, e.g., touch tone navigation systems.

The customers 110 may communicate to the central offices 120 usingvoice, telephone, fax, or data accessories (not shown). When connectingto the telephony system, the telephones or other telephony equipmentgenerate sinusoids at specified frequencies according to the DTMF,MF-R1, or MF-R2 protocol used to encode the dialed digits.

Most computer users are familiar with dialed digit tones produced by acomputer modem when dialing to a service provider and the dialed digittones produced by a telephone speaker corresponding to the telephonekeypad digits. The dialed digit tones produced and heard comprise twoaudible frequencies resulting from the speaker acting as a transducer,converting electrical sinusoids to the audible tones. It is theseelectrical sinusoids that are detected and decoded as dialed digits byequipment in a central office 120. Before discussing details of anembodiment of the equipment employed in a central office 120, a generaldescription of an embodiment employed by a media gateway is presented inFIG. 1B.

FIG. 1B is a block diagram of a media gateway 150 comprising an echocanceler 175, detector 180, and vocoder (i.e., voice-coder) 185. In thisexample, the media gateway 150 is shown as a network device facilitatingvoice-over-Internet Protocol (VOIP). Voice-over-IP is an inexpensivemeans for communicating across long distances. A voice signal isconverted into network packets by the media gateway 150. The networkdevice receiving the IP packet 190 containing the voice signalinformation converts the data in the network packet 190 back into avoice signal. Radio stations and other broadcast facilities may employmedia gateways 150 to broadcast signals over the Internet 160 (FIG. 1A).

The echo canceler 175 removes echoes caused by leakage of electricalsignals from hybrids (not shown), having an impedance mismatch, in themedia gateway 150 and central office 120, for example. The detector 180determines sinusoidal tones, indicating dialed digits, received from thecentral office 120. The detector 180 can be configured to detect DTMFtones, MF-R1 tones, or MF-R2 tones. The vocoder 185 is employed tocompress speech signals to increase the bandwidth efficiency. Finally, apacketizer 187 converts the vocoder output to IP packets 190, or otherforms of data packets used to transmit data across the data network 100.The IP packets 190 are transmitted across the IP network 160 to anaddress corresponding to the dialed digits detected by the detector 180.It should be understood that the vocoder 185 can be other forms ofencoders, such as video encoders to transmit video streams rather thanaudio streams. The echo canceler may be of the type described in U.S.patent application Ser. No. 09/350,497, incorporated herein by referencein its entirety.

FIG. 2 is a block diagram of equipment in the central office 120 that isused to detect sinusoids used to encode the dialed digits. The circuitboards 200 comprise an analog-to-digital (A/D) converter 210, digitalsignal processor (DSP) 220, and external memory 230.

The A/D converter 210 receives an analog signal 205, which is acontinuous-time form of an electrical signal, from telephony equipment(not shown) used by the customer 110. The analog signal 205 comprisesthe sinusoids used to encode the dialed digits. The A/D converter 210converts the analog signal 205 to a corresponding digital signal 215 ata sampling rate of about 4 kHz.

The DSP 220 comprises internal memory 250, which may include cache, RAM,or ROM, i.e., volatile or non-volatile memory components. The DSPfurther comprises detectors 240 a-240 d (collectively 240). Thedetectors 240, in the DSP 220 case, are software programs executed bythe DSP 220. Because the detectors 240 are implemented in software, theycan be quickly changed among DTMF, MF-R1, and MF-R2 detectors to supportthe dialed digit protocol of the calling device (not shown).

During execution, the detectors 240 access the internal memory 250 forvarious typical reasons, such as recalling multiplier coefficients orstoring intermediate arithmetic results. The detectors 240 may alsoaccess external memory 230 for similar or other reasons. Becauseaccessing the external memory 230 typically takes more time thanaccessing the internal memory 250, initialization parameters for thedetectors 240 are generally stored in the external memory 230, while theDSP 220 typically employs the internal memory 250 for real-time memoryusages.

FIG. 3 is an embodiment of a DTMF detector 240 a executed by the DSP 220of FIG. 2. The DTMF detector 240 a comprises: a PS-IIR band-split filter300, banks of filters referred to as (i) DTMF low-band notch filters 810a and (ii) DTMF high-band notch filters 810 b, respectively, and DTMFdetection logic 850 a.

The PS-IIR band-split filter 300 receives the digital signal 215 fromthe A/D converter 210 (FIG. 2). The PS-IIR band-split filter 300subdivides the digital signal 215 into a 2 kHz low-band signal 350,comprising frequencies between 0 Hz and about 1 kHz, and a 2 kHzhigh-band signal 360, comprising frequencies between about 1 kHz and 2kHz. Because the 2 kHz low-band signal comprises frequencies between 0Hz and about 1 kHz, the 2 kHz low-band signal 350 is sometimes referredto as a 0-1 kHz subband. Also, because the 2 kHz high-band signal 360comprises frequencies between about 1-2 kHz, the 2 kHz high-band signal360 is sometimes referred to as a 1-2 kHz subband. The low-band signal350 and high-band signal 360 may also be referred to as simply E₀ andE₁, respectively, where “E” denotes “energy.”

In the DTMF detector 240 a, E₀ is processed by the DTMF low-band notchfilters 810 a, and E₁ is processed by the DTMF high-band notch filters810 b. The DTMF detection logic 850 a processes the outputs of each ofthe banks of notch filters 810 a, 810 b. Both the banks of filters 810a, 810 b and the detection logic 850 a demand less processing time thansimilar structures processing signals comprising frequency components upto 4 kHz, which is why the PS-IIR band split filter 300 is employed.

FIG. 4 is a block diagram of the PS-IIR band-split filter 300. ThePS-IIR band-split filter 300 is depicted in polyphase form. An inputswitch 305 alternates the input between a zero'th first-order all-passsection 310 and a first first-order all-pass section 320. Outputs fromfirst-order all-pass sections 310, 320 are added by an adder 330,producing the 2 kHz low-band signal 350; outputs from the first-orderall-pass sections 310, 320 are subtracted by a subtractor 340, producingthe 2k high-band signal 360.

PS-IIR filters have the following transfer function $\begin{matrix}{{{H_{L}(z)} = {\frac{1}{2}\left( {{A_{o}\left( z^{2} \right)} + {z^{- 1}{A_{1}\left( z^{2} \right)}}} \right)}},{where}} & (1) \\{{{A_{i}\left( z^{2} \right)} = {\prod\limits_{j = o}^{p_{i} - 1}\frac{\alpha_{i,j} + z^{- 2}}{1 + {\alpha_{i,j}z^{- 2}}}}},\quad{i = 0},1} & (2)\end{matrix}$

are cascaded, second-order, all-pass sections. Various design methodsexist to find the optimal set of α_(i,j) to realize a desired frequencyresponse. For more information regarding finding optimal sets ofα_(i,j), see the following references: P. P. Vaidyanathan, Multi-rateSystems and Filter Banks, Prentice-Hall, 1993; O. Tanrikulu and M.Kalkan, “Design and Discrete Re-optimization of All-pass Based PowerSymmetric IIR Filters,” Electronics Letters, vol. 32, no. 16, pp.1458-1460, 1996; R. A. Valenzuela and A. G. Constantinides, “DigitalSignal Processing Schemes for Efficient Interpolation and Decimation,”IEEE Proceedings-G, Circuits Devices and Systems, vol. 130, no. 6,pp.225-235, 1983; O. Tanrikulu, Adaptive Algorithms for AcceleratedConvergence and Noise Immunity, Ph.D. Thesis, Imperial College ofScience, Technology and Medicine, 1995.

For the problem of detecting dialed digits encoded by sinusoids in anelectrical signal, a PS-IIR filter splits the incoming electrical signalinto two parallel signal paths at half (i.e., 2 kHz) the input samplingfrequency (i.e., 4 kHz). Splitting the incoming signal into two parallelsignal paths at half the input sampling frequency is desired to reducethe complexity of the detector and improve detector robustness.

The parallel signal paths separate the row and column frequencies of theDTMF case. Here, H₁(z) is a low-pass filter. Thus, to separate the rowand column frequencies, a mirror-image high-pass filter is needed, so,by using a transformation z→−z, the mirror-image high-pass filter is:$\begin{matrix}{{H_{H}(z)} = {\frac{1}{2}{\left( {{A_{o}\left( z^{2} \right)} - {z^{- 1}{A_{j}\left( z^{2} \right)}}} \right).}}} & (3)\end{matrix}$

Since H_(L)(z) and H_(H)(z) are structurally similar, the subbandsplitting operation can be implemented through the polyphaserepresentation of FIG. 4. Note that A_(i)(z²)→A_(i)(z) becomes a cascadeof first-order all-pass sections. Furthermore, the filtering operationsare conducted at the lower sampling rate. And, there is still room forimprovement in the implementation of A_(i)(z) by using a compactrealization of all-pass transfer functions rather than a canonicalrealization. See S. K. Mitra and K. Hirano, “Digital All-pass Networks,”IEEE Trans. Circuits Systs., vol. CAS-21, no. 5, pp. 688-700, September1974.

FIG. 5 is a block diagram of a compact implementation of a first-orderall-pass section 400. The first-order all-pass section 400 comprisesadders 330, subtractor 340, unit delay 410, and multiplier 420. The unitdelay 410 is sometimes referred to as a storage element. The multiplier420 multiplies a signal by a coefficient, α_(i,j).

An advantage of the implementation in FIG. 5 over less compactimplementations is that the first-order all-pass section 400 has onlyone storage element and a single multiplier. Other implementations thattrade storage against the number of additions and multiplications canalso be used. See S. K. Mitra and K. Hirano, “Digital All-passNetworks,” IEEE Trans. Circuits Systs., vol. CAS-21, no. 5, pp. 688-700,September 1974. Finally, this implementation is structurally passive andlends itself well to programming on a fixed-point DSP. See O. Tanrikulu,B. Baykal, A. G. Constantinides, J. A. Chambers and P. A. Naylor,“Finite-precision design and implementation of all-pass polyphasenetworks for echo cancellation in sub-bands,” ICASSP-95, Detroit, USA,vol., 4, pp. 3039-3042, May 1995.

A cascade of two first-order all-pass sections 400 is employed in eachsignal path of the PS-IIR band-split filter 300. In other words, boththe zero'th first-order all-pass section 310 and first first-orderall-pass section 320 comprises two first-order all-pass sections 400, asdefined by equation (2) and depicted in FIG. 5.

Amplitude spectra of H_(L)(z) and H_(H)(z) and the frequency spectrallines composing the DTMF, MF-R1 and MF-R2 protocols are shown togetherin FIGS. 6-8, respectively. Seeing the relationships among filterfrequencies and sinusoid frequencies is useful for visualizing how theDTMF detector 240 a works in detecting the dialed digits.

FIG. 6 is a plot comprising transfer functions of the band-split filter300 (FIG. 3) and spectral lines of the frequencies defined in the DTMFprotocol. A low-pass filter transfer function 510 is a frequencyresponse of the low pass filter, H_(L)(z) of equation (1), from thePS-IIR band-split filter (FIG. 4). A high-pass filter transfer function520 is a frequency response of the high pass filter, H_(H)(z) ofequation (3), from the PS-IIR band-split filter (FIG. 4). The spectrallines 530 correspond to DTMF row/column frequencies: row frequencies 697Hz, 770 Hz, 852 Hz, 941 Hz; column frequencies 1209 Hz, 1336 Hz, 1477Hz, and 1633 Hz.

As suggested by the relationships of the transfer functions 510, 520 andthe spectral lines 530 of the DTMF frequencies, the PS-IIR band-splitfilter 300 sufficiently isolates the row frequencies from the columnfrequencies. Here, the PS-IIR band split filter is designed to isolatethe row and column frequencies at 1 kHz, which is determined by the −3dBpoint on the transfer functions 510, 520 corresponding to each of thelow- and high-pass filters comprising the PS-IIR filter. However, therow and column frequencies are not symmetrically distributed withrespect to the band-split filters. Therefore, since the band-splitfilters are not perfect (i.e., finite stop-band attenuation) incombination with decimation of the 4 kHz input signal at 2 kHz (i.e.,the input switch 305, FIG. 3), leakage of the energy of the rowfrequencies adds to the energy observed as column frequencies, andvice-versa. This has an impact on the detector design, which is found inthe scale factors, ∝_(r) and ∝_(c), having different values.

In summary, the 2 kHz low-band signal 350 (FIG. 3) comprises the rowfrequencies, namely 697 Hz, 770 Hz, 852 Hz and 941 Hz that are extractedfrom the digital signal 215 by the low-pass filter of the PS-IIR filter300. The 2 kHz high-pass signal 360 (FIG. 3) comprises the columnfrequencies, namely 1209 Hz, 1336 Hz, 1477 Hz, and 1633 Hz, that areextracted from the digital signal 215 by the high-pass filter of thePS-IIR band split filter 300. The resulting 2 kHz subband signals areprocessed separately where advantageous to do so.

Referring again to FIG. 3, the banks of filters 810 a, 810 b comprisenotch filters at row and column frequencies within the respectivesubbands 350, 360. Outputs from the banks of filters 810 a, 810 b arereceived by the DTMF detection logic 850 a.

FIG. 9 comprises the DTMF detection logic 850 a (FIG. 3). The detectorstarts in step 905 upon receiving the digital signal 215 (FIG. 3). ThePS-IIR band split filter 300 separates the digital signal 215 intoenergy signals E₀, E₁, as described above. The PS-IIR band split filter300 has a property that the sum of the energies, E₀+E₁, is equal to theenergy of the digital signal 215. Thus, a first check is made todetermine whether the requisite amount of energy is found in the digitalsignal 215 received by the DTMF detector 240 a (FIG. 3). A signal powerlevel comparator 910 performs the comparison:E₀+E₀>?−40 dBm0.  (4)

If the sum of the energies of E₀ and E₁ are not greater than −40 dBm0,the notch filters do not operate on the incoming signals, no DTMF isdeclared, and processing terminates in step 915. Next, a DTMF twist test920 a is performed, where|E₀−E₁|>?6 dB  (5)

If the twist test 920 a fails, DTMF is not declared, and processingterminates in step 915. Note that this check also prevents speechactivity from being falsely detected as DTMF later on, thereby reducingerrors due to talk-offs. If the twist test 920 a does not fail, thenprocessing continues with the execution of the DTMF low-band notchfilters 810 a.

The notch filter with the lowest output energy is (i) identified at theoutput of the DTMF low-band notch filters 810 a in step 925 a and (ii)compared in step 930 to the corresponding input E₀ using the comparisonformula E₀>?E_(0,c)α_(c). This comparison yields whether significantenergy is residing in the respective notch bandwidth.

If the result of the comparison is negative, DTMF is not declared andprocessing terminates in step 915. Thus, the processing for the DTMFhigh-band notch filter 810 b is skipped altogether. Either the signaldid not satisfy the bandwidth requirements or it was speech and atalk-off signal is presented. If a valid column frequency is detected bythe column energy comparitor 930, however, the above procedure isrepeated for the DTMF high-band notch filters 810 b to determine whetherE_(i) comprises a valid row frequency.

A row minimum selector 935 selects the notch filter with the lowestoutput energy. The energy of the output of the selected notch filter iscompared to the corresponding input E₁. This comparison yields whethersignificant energy was residing in that particular notch bandwidth.Similar to the column energy comparator 930, if the row energycomparitor 945 determines that the input energy, E₁, is greater thanE_(1,r) multiplied by the scale factor ∝_(r) (i.e., E₁>E_(i,r)∝_(r)),then a valid row frequency is not declared, and the process terminatesin step 915. Otherwise, in other words, if valid row and columnfrequencies are detected, then a valid DTMF is declared for thatparticular frame of data. As a last protection from false detections, itmay be required to detect the DTMF for a number of frames of data todeclare a DTMF is received. Note that the process of FIG. 9 is for asingle frame of input data.

It should be noted, and it is also true in MF-R1 and MF-R2, that thethresholds used for input/output energy comparisons are not the same forE₀ and E₁. The reason is that, as observable in FIGS. 6-8, thefrequencies of interest and the subband split are not symmetrical withrespect to 1000 Hz.

MF-R1 Detection

FIG. 10 is a block diagram for MF-R1 detection. An MF-R1 detector 240 bis similar to the DTMF detector 240 a (FIG. 3) with two differences.First, the 2 kHz low-band signal 350 contains only two frequencies ofinterest, which can be observed in the frequency plot of FIG. 7.

Referring to FIG. 7, it is observed that the low-pass filter transferfunction 510 and the high-pass filter transfer function 520 describingrespective filters composing the PS-IIR band split filter 300 are thesame as those for the DTMF detector 240 a. However, the MF-R1frequencies 540 fall within the unity gain regions of the two filters510, 520 in an unequal manner. Specifically, frequencies 700 Hz and 900Hz are passed by the low-pass filter of the PS-IIR band-split filter300, while frequencies 1100 Hz, 1300 Hz, 1500 Hz, and 1700 Hz, arepassed by the high-pass filter of the PS-IIR band-split filter 300.Sufficient rejection is provided against frequencies not within therespective passbands of the low and high pass filters.

In the MF-R1 protocol, there is no row/column frequency, as in the DTMFprotocol. So, when there is an MF-R1 digit, both frequencies can belocated in the same subband. Therefore, the detection logic 850 b forthe MF-R1 signals (i.e., sinusoids) is different from the detectionlogic 850 a (FIG. 3) for the DTMF signals.

FIG. 11A is a flow diagram of an MF (MF-R1/R2) detector 850 b employedby the MF-R1 detection logic. The MF detector 850 b starts in step 905upon receiving the digital signal 215. The first step of the MF detector850 b, i.e. absolute energy comparison (step 910) , is identical to step910 of the DTMF detector 850 a. In the absolute energy comparison, thelower and upper subband energies E0 and E1 are added together andcompared with the absolute threshold (−40 dbm0). If the total energy isgreater than −40 dBm0, the MF detector 850 b proceeds to the next step,the Twist test (step 933). Here, unlike the DTMF case, there are threepossible outcomes, referred to as Case 1, Case 2, and Case 3. Case  1:$\quad{0.14 \leq \frac{E_{0}}{E_{1}} \leq 7.08}$

Case 1 corresponds to the upper and lower subband energies being within+/−8.5 dB of each other. In this case, there is a high likelihood thattones are present in both the upper and lower subbands. The MF detector850 b then proceeds exactly as in the case of DTMF, picking one tonefrom each subband. The bandwidth test is then performed using the lowestoutput energies from the upper and lower subband notches, as determinedin steps 925 c, 925 b, respectively. If E₀>E_(0,1)α₁ (step 945 a) andE₁>E_(1,u)α_(u) (step 930 a), where E_(0,1) and E_(1,u) are the minimumoutput energies for the lower and upper subbands respectively, then, instep 950 a, the MF detector 850 b picks the digit referenced by the twotones. Case  2: $\quad{\frac{E_{0}}{E_{1}} > 7.08}$

FIG. 11B is a flow diagram of an embodiment of a process of Case 2 ofthe MF detector. In Case 2, it may be that either (i) there is a twistlarger then +8.5 dB in the MF signal (with tones present in bothsubbands) or (ii) all the energy is present in the upper subband. Case 2then searches for the lowest two output energies, E_(o,u1) (step 925 d)and E_(o,u2) (step 925 e) from the four upper subband frequencies.

The first test (step 936 a) checks to see how noisy the signal is. If${\frac{E_{0,{u\quad 1}} + E_{o,{u\quad 2}}}{E_{0}} < 1.35},$the signal is assumed to be relatively noise free. The rationale behindthe test of step 936 a is that as the noise level in the signal rises(noise in this case being any signal that is not an MF frequency tone),the ratio of $\frac{E_{0,{u\quad 1}} + E_{o,{u\quad 2}}}{E_{0}}$gets closer and closer to 2.00. For a white noise signal, and usingnotches of infinitesimal width, the ratio approaches 2.00. On the otherhand, with no background noise (i.e. only tonal energy), the ratioapproaches 1.00. The test of step 936 a is useful for improving thetalk-off performance of the MF detector 850 b.

The next test (step 939 a) checks to see if there are really twodistinct tones present in the digital signal 215. If${0.1 < \frac{E_{0,{u\quad 1}}}{E_{0,{u\quad 2}}} < 10.0},$it is likely that there are two tones in the signal, since the outputenergies from either of the two notches are good estimators of theenergy in the other tone in the subband, given that the signal isrelatively noise free.

The last test (steps 945 b, 945 c) performed is the bandwidth test. IfE₀>E_(o,u1)α_(u) and E_(o)>E_(o,u2)α_(u), then in step 950 b, the MFdetector 850 b picks the digits referenced by the two tones. Case  3:$\quad{\frac{E_{0}}{E_{1}} < 0.14}$

FIG. 11C is a flow diagram of an embodiment of a process of Case 3 ofthe MF detector 215. Case 3 is the last case in the MF detector logic.Again, it can mean that either all the MF energy is in the lower subbandor the twist is greater than −8.5 dB. Since there are only two possibleMF frequencies in the lower subband, no search needs to be performed.The output energies E_(1,10) and E_(1,11) are used to carry out a noisysignal test (step 936 b), single tone rejection test (step 939 b), andbandwidth test (steps 930 b, 930 c) (as in case 2 above). If all thetests pass, the MF detector 850 b picks MF digit “1” in step 950 c.

MF-R2 Detection

Two detectors for forward and backward frequencies are necessary in theMF-R2 protocol case. The MF-R2 forward frequency detector 240 c isdepicted in FIG. 12. The MF-R2 backward frequency detector 240 d isdepicted in FIG. 13.

Referring to FIG. 12, the forward frequencies do not include frequenciesbelow 1 kHz. Therefore, there is no 2 kHz low-band signal 350. The 2 kHzhigh-band signal 360 comprises six signals, which are filtered by a bankof six notch filters 830 a. The forward frequencies are indicated inFIG. 8 as the vertical dashed frequency lines 550. The six MF-R2 forwardfrequencies, namely 1380 Hz, 1500 Hz, 1620 Hz, 1740 Hz, 1860 Hz, and1980 Hz are passed by the high-pass filter of the PS-IIR band splitfilter 300 (FIG. 12), as represented by the high-pass filter transferfunction 520.

FIG. 13 is a block diagram of an MF-R2 backward detector 240 d. TheMF-R2 protocol comprises sinusoids at backward frequencies below 1 kHzand above 1 kHz, so there are two banks of notch filters 830 b, 830 c,respectively, to filter corresponding sinusoids. The MF-R2 backwarddetector 240 d comprises an extra notch filter 840 to remove 980 Hz fromthe lower subband.

Referring again to FIG. 8, the MF-R2 backward frequencies 560 are solidfrequency spectra lines in the frequency plot. The backward frequenciesare 540 Hz, 660 Hz, 780 Hz, 900 Hz, 1020 Hz, and 1140 Hz. The backwardfrequency 1020 Hz is within the transition band of both thelow-frequency filter and high-frequency filters of the PS-IIR filter 300(FIG. 13). Therefore, this frequency shows up in both subbands, E₀ andE₁. To avoid an extra signal from causing an error, the notch filter 840is used to remove the 1020 Hz frequency from the lower subband.

The DTMF detector 240 a passes the Net-4 European tests with nofailures. Talk-off tests for the entire Bell-Core test signals yieldonly 5 talk-off cases compared to an existing DTMF detector, which isone-fifth the density and fails around 250 times. Existing systemsprovide approximately 300 channels per circuit board, whereas a systememploying the filters, etc., described above provides 1500 channels.Through another level of optimization, 2500 channels per board arepossible, as described below in reference to FIGS. 14-15.

Since there are eight notch filters but only two signals being detected,there are six notch filters needlessly executing instructions, thuswasting execution cycles. Saving the execution cycles allows for anincrease in the number of detectors that may be executed by the DSP 220(FIG. 2). To determine which two notch filters are necessary fordetecting sinusoids present in the digital signal 215 corresponding tothe analog signal 205, a preclassifier may be employed. Thepreclassifier performs frequency estimation to select which two notchfilters should be used. Though requiring execution cycles, thepreclassifier saves six of eight notch filters from wasting executioncycles, thus increasing the number of detectors per board from 1500 to2500.

FIG. 14 is a modified DTMF detector 240 e. The modified DTMF detector240 e comprises preclassifiers 805 a and 805 b. The preclassifiers 805a, 805 b receive respective subbands 350, 360 from the PS-IIR band splitfilter 300. Each preclassifier 805 a, 805 b performs frequencyestimation to determine which notch filter(s) within the respectivebanks of notch filters 810 a, 810 b should be executed to determinewhether the DTMF signals are present.

FIG. 15 is a flow diagram of an embodiment of a modified DTMF detectionlogic 850 c employed by the modified DTMF detector 240 e. The modifiedDTMF detection logic 850 c comprises the preclassifiers 805 a, 805 b andindicates their relative positions within the process. Other than thepreclassifiers 805 a, 805 b, the modified DTMF detection logic 850 c isthe same as the DTMF detection logic 850 a described in FIG. 9.

The signaling detectors described herein take advantage of subbanddecomposition using IIR filter-banks. This brings high computationalefficiency and low memory costs which are useful in high densityapplications, such as DTMF detection, and increases the number ofchannels, or customers, a service provider can support from a centraloffice.

Equivalents:

The principles of the present invention allow for the embodimentsdescribed herein to be expanded to other forms of encoding protocols,such as protocols comprising three, four, or more sinusoids. Band-rejectand band-pass filters rather than strictly low-pass and high-passfilters to produce corresponding three, four, or more subdivisions maybe employed; a similar set of filters may be employed in an alternativeembodiment of the PS-IIR band-split filter 300 (FIG. 3).

The detection logic, filters, and other aspects of the processes andfunctions described herein are not restricted to any particular softwarelanguage or data structure. The hardware depicted is merely exemplary.Alternative processors, from analog circuits to ASICs (applicationspecific integrated circuits), may be employed.

Although subband filtering is employed to dissect the time domain signalby determining a frequency domain signal equivalent, alternativetransforms yielding a one-to-one mapping between the time and frequencydomains may be employed. Examples of alternative transforms include: DFT(discrete Fourier transform), DHT (discrete Hartley transform), DCT(discrete Cosine trasform), Wavelets, etc. Typical processing followingthe alternative transforms may vary according to the respectivetransforms but are still within the scope of the principles of thepresent invention.

While this invention has been particularly shown and described withreferences to preferred embodiments thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the scope of the inventionencompassed by the appended claims.

1. A method of determining a presence of sinusoids in an electricalsignal, comprising: splitting the electrical signal into subbands havinga common sampling frequency of less than twice the highest frequency ofsinusoids in the electrical signal; and analyzing energies within thesubbands at the common sampling frequency to determine the presence ofthe sinusoids.
 2. The method according to claim 1, wherein the sinusoidscorrespond to frequencies of dialed digits.
 3. The method according toclaim 1, wherein splitting the electrical signal into the subbandscomprises extracting subbands of 0-1 kHz and 1-2 kHz.
 4. The methodaccording to claim 1, wherein splitting the electrical signal into thesubbands comprises filtering the electrical signal using a powersymmetric infinite impulse response (PS-IIR) filter.
 5. The methodaccording to claim 4, wherein the PS-IIR filters are implemented in apolyphase form.
 6. The method according to claim 4, wherein the PS-IIRfilters comprise all-pass sections implemented in compact realizations.7. The method according to claim 1, further comprising filtering thesubbands with at least one bank of filters comprising filterscorresponding to the number of possible frequencies of the sinusoidswithin the respective subbands.
 8. The method according to claim 7,wherein the filters are notch filters.
 9. The method according to claim7, wherein, for DTMF detection, splitting the electrical signalcomprises (i) extracting a 0-1 kHz subband and a 1-2 kHz subband and(ii) filtering the subbands with four notch filters per bank of filters.10. The method according to claim 7, wherein, for MF-R1 detection,splitting the electrical signal comprises (i) extracting a 0-1 kHzsubband and a 1-2 kHz subband and (ii) filtering the 0-1 kHz subbandwith two notch filters and the 1-2 kHz subband with four notch filters.11. The method according to claim 7, wherein: for MF-R2 forwarddetection, splitting the electrical signal comprises (i) extracting a0-1 kHz subband and 1-2 kHz subband and (ii) filtering the 1-2 kHzsubband with six notch filters; and for backward detection, splittingthe electrical signal comprises (i) extracting a 0-1 kHz subband and a1-2 kHz subband and (ii) filtering the 0-1 kHz subband with a notchfilter at 980 Hz, to remove aliasing of a 1020 Hz tone in the 1-2 kHzsubband, and four other notch filters and the 1-2 kHz subband with twonotch filters.
 12. The method according to claim 7, further comprisingpreclassifying the sinusoids in the subbands and selecting filterswithin respective banks of filters that match frequencies of thepreclassified sinusoids.
 13. The method according to claim 1, whereinanalyzing energies comprises determining whether a summing of theenergies in the subbands exceeds a minimum threshold level.
 14. Themethod according to claim 1, wherein analyzing energies comprisesdetermining whether a difference between the energies in the subbands isbelow a twist-test threshold.
 15. The method according to claim 1,wherein for each subband, analyzing energies comprises comparing energylevels of an output of a notch filter having a lowest output energylevel from among at least two notch filters in a bank of filters to theenergy of the input signal to the bank of filters.
 16. The methodaccording to claim 1, wherein analyzing energies further comprisesreporting valid dialed digits.
 17. The method according to claim 1,wherein the electrical signal is sampled by an analog-to-digitalconverter and splitting the electrical signal and analyzing energies isexecuted by a digital processor.
 18. An apparatus, comprising: asplitter to separate an electrical signal into subbands having a commonsampling frequency of less than twice the highest frequency of sinusoidsin the electrical signal; and an analyzer to measure energies within thesubbands at the common sampling frequency to determine a presence of thesinusoids.
 19. The apparatus according to claim 18, wherein thesinusoids correspond to frequencies of dialed digits.
 20. The apparatusaccording to claim 18, wherein the splitter extracts subbands of 0-1 kHzand 1-2 kHz.
 21. The apparatus according to claim 18, wherein thesplitter comprises a power symmetric infinite impulse response (PS-IIR)filter to separate the electrical signal into the subbands.
 22. Theapparatus according to claim 18, further comprising at least one bank offilters to filter the subbands, the bank of filters comprising filterscorresponding to the number of possible frequencies of sinusoids withinthe respective subbands.
 23. The apparatus according to claim 22,wherein the filters are notch filters.
 24. The apparatus according toclaim 22, further comprising at least one preclassifier to determine thesinusoids in the subbands and to select filters within respective banksof filters that match frequencies of the sinusoids.
 25. The apparatusaccording to claim 18, wherein the electrical signal is sampled by ananalog-to-digital converter and the splitter and analyzer areimplemented in digital processor instructions and executed by a digitalprocessor.
 26. The apparatus according to clam 18, being employed in adevice supporting voice-over-IP.
 27. An apparatus, comprising: ananalog-to-digital converter to sample a received analog signal and tooutput a corresponding digital signal; and a digital processor coupledto an output of the analog-to-digital converter to receive the digitalsignal and to execute program instructions to: split the digital signalinto subbands having a common sampling frequency of less than twice thehighest frequency of sinusoids in the digital signal; and analyzeenergies within the subbands at the common sampling frequency todetermine a presence of the sinusoids.
 28. A computer-readable mediumhaving stored thereon sequences of instructions, the sequences ofinstructions including instructions, when executed by a processor,causes the processor to perform: splitting an electrical signal intosubbands having a common sampling frequency of less than twice thehighest frequency of sinusoids in the electrical signal; and analyzingenergies within the subbands at the common sampling frequency todetermine a presence of sinusoids.
 29. A voice-over-IP device,comprising: a receiver to receive electrical signals composed of voicesignals and dialed digit sinusoids corresponding to dialed digits; and adetector coupled to the receiver to monitor the electrical signals andto detect the dialed digit sinusoids, said detector including: asplitter to split the electrical signal into subbands having a commonsampling frequency of less than twice the highest frequency of thedialed digit sinusoids; and an analyzer to analyze energies within thesubbands at the common sampling frequency to determine a presence of thesinusoids; and a generator to generate packets of data comprising (i)voice signal data and (ii) information corresponding to the dialeddigits.